rev. 0 information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a ad8614/ad8644 one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700 world wide web site: http://www.analog.com fax: 781/326-8703 ? analog devices, inc., 1999 single and quad +18 v operational amplifiers pin configurations 5-lead sot-23 (rt suffix) 1 2 3 5 4 2 in +in v+ out a ad8614 v 2 14-lead tssop (ru suffix) out a 2 in a 1 in a v 1 2 in d 1 in d v 2 out d 114 1 in b 2 in b out b 2 in c out c 1 in c 78 ad8644 14-lead narrow body so (r suffix) 14 13 12 11 10 9 8 1 2 3 4 5 6 7 Cin a +in a v+ +in b Cin b out b out d Cin d +in d vC +in c Cin c out c out a ad8644 features unity gain bandwidth: 5.5 mhz low voltage offset: 1.0 mv slew rate: 7.5 v/ m s single-supply operation: 5 v to 18 v high output current: 70 ma low supply current: 800 m a/amplifier stable with large capacitive loads rail-to-rail inputs and outputs applications lcd gamma and v com drivers modems portable instrumentation direct access arrangement general description the ad8614 (single) and ad8644 (quad) are single-supply, 5.5 mhz bandwidth, rail-to-rail amplifiers optimized for lcd monitor applications. they are processed using analog devices high voltage, high speed, complementary bipolar processhv xfcb. this proprietary process includes trench isolated transistors that lower internal parasitic capacitance which improves gain bandwidth, phase mar- gin and capacitive load drive. the low supply current of 800 m a (typ) per amplifier is critical for portable or densely packed designs. in addition, the rail-to-rail output swing provides greater dynamic range and control than standard video amplifiers provide. these products operate from supplies of 5 v to as high as 18 v. the unique combination of an output drive of 70 ma, high slew rates, and high capacitive drive capability makes the ad8614/ad8644 an ideal choice for lcd applications. the ad8614 and ad8644 are specified over the temperature range of C20 c to +85 c. they are available in 5-lead sot-23, 14-lead tssop and 14-lead soic surface mount packages in tape and reel.
C2C rev. 0 ad8614/ad8644Cspecifications electrical characteristics parameter symbol conditions min typ max unit input characteristics offset voltage v os 1.0 2.5 mv C20 c t a +85 c3mv input bias current i b 80 400 na C20 c t a +85 c 500 na input offset current i os 5 100 na C20 c t a +85 c 200 na input voltage range 0v s v common-mode rejection ratio cmrr v cm = 0 v to v s 60 75 db voltage gain a vo v out = 0.5 v to v s C0.5 v, r l = 10 k w 10 150 v/mv output characteristics output voltage high v oh i load = 10 ma v s C0.15 v output voltage low v ol i load = 10 ma 65 150 mv output short circuit current i sc 35 70 ma C20 c t a +85 c30 ma power supply psrr psrr v s = 2.25 v to 9.25 v 80 110 db supply current / amplifier isy 0.8 1.1 ma C20 c t a +85 c 1.5 ma dynamic performance slew rate sr c l = 200 pf 7.5 v/ m s gain bandwidth product gbp 5.5 mhz phase margin f o 65 degrees settling time t s 0.01%, 10 v step 3 m s noise performance voltage noise density e n f = 1 khz 12 nv/ ? hz e n f = 10 khz 11 nv/ ? hz current noise density i n f = 10 khz 1 pa/ ? hz note all typical values are for v s = 18 v. specifications subject to change without notice. (5 v v s 18 v, v cm = v s /2, t a = 25 8 c unless otherwise noted) caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the ad8614/ad8644 features proprietary esd protection circuitry, permanent dam- age may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. warning! esd sensitive device absolute maximum ratings 1 supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 v input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . gnd to v s storage temperature range . . . . . . . . . . . . C65 c to +150 c operating temperature range . . . . . . . . . . . C20 c to +85 c junction temperature range . . . . . . . . . . . . C65 c to +150 c lead temperature range (soldering, 60 sec) . . . . . . . . 300 c notes 1 stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. exposure to absolute maximum rating condi- tions for extended periods may affect device reliability. package type u ja 1 u jc unit 5-lead sot-23 (rt) 230 140 c/w 14-lead tssop (ru) 180 35 c/w 14-lead soic (r) 120 56 c/w note 1 q ja is specified for worst-case conditions, i.e., q ja is specified for device soldered onto a circuit board for surface mount packages. ordering guide temperature package package model range description option ad8614art 1 C20 c to +85 c 5-lead sot-23 rt-5 ad8644aru 2 C20 c to +85 c 14-lead tssop ru-14 ad8644ar 2 C20 c to +85 c 14-lead soic r-14 notes 1 available in 3,000 or 10,000 piece reels. 2 available in 2,500 piece reels only.
ad8614/ad8644 C3C rev. 0 typical performance characteristics C capacitance C pf 10 10k 100 1k 50 45 0 40 35 30 25 20 15 10 5 v s = 18v r l = 2k v t a = 25 8 c +os 2 os small signal overshoot C % figure 1. small signal overshoot vs. load capacitance settling time C m s output swing from 0 to 6 v 12 2 12 0 0.5 3.5 1.0 1.5 2.0 2.5 3.0 8 4 0 2 4 2 8 0.1% 0.01% 0.1% 0.01% figure 2. settling time gain C db frequency C hz 1k 100m 10k 100k 1m 10m 80 60 40 20 0 45 90 135 180 5v # v s # 18v r l = 1m v c l = 40pf t a = 25 8 c phase shift C degrees figure 3. open-loop gain and phase vs. frequency time C 1 m s/div v s = 5v r l = 2k v c l = 200pf a v = 1 t a = 25 8 c voltage C 1v/div 7.5 6.5 5.5 4.5 3.5 2.5 1.5 0.5 2 0.5 2 1.5 2 2.5 figure 4. large signal transient response time C 1 m s/div v s = 18v r l = 2k v c l = 200pf a v = 1 t a = 25 8 c voltage C 4v/div 29 25 21 17 13 9 5 1 2 3 2 7 2 11 figure 5. large signal transient response time C 500ns/div v s = 5v # v s # 18v r l = 2k v c l = 200pf a v = 1 t a = 25 8 c v s 2 voltage C 50mv/div figure 6. small signal transient response load current C ma 10 1 0.001 100 0.01 0.1 1 10 100 1k source sink 5v # v s # 18v t a = 25 8 c 10k d output voltage C mv figure 7. output voltage to supply rail vs. load current supply voltage C 6 volts 1,000 400 0 010 12345 6789 900 500 300 100 700 600 200 800 t a = 25 8 c supply current/amplifier C m a figure 8. supply current vs. supply voltage common-mode voltage C volts 400 0 2 400 2 2.5 2.5 2 1.5 2 0.5 0.5 1.5 300 200 2 200 2 300 100 2 100 v s = 6 2.5v input bias current C na figure 9. input bias current vs. common-mode voltage
ad8614/ad8644 C4C rev. 0 common-mode voltage C volts 400 0 2 400 2 9 9 2 7 2 5 2 3 2 1 01 35 7 300 200 2 200 2 300 100 2 100 v s = 6 9v input bias current C na figure 10. input bias current vs. common-mode voltage input offset voltage C mv 2 2 2 0 0.5 1 1.5 quantity C amplifiers 180 160 0 80 60 40 20 140 100 120 2 1.5 2 1 2 0.5 2.5v # v s # 9v t a = 25 8 c figure 11. input offset voltage distribution temperature C 8 c supply current/amplifier C ma 1.0 0.9 0.5 2 35 2 15 5 25456585 0.8 0.7 0.6 v s = 18v v s = 5v figure 12. supply current vs. temperature frequency C hz output swing C v p-p 6 5 0 100 1k 10m 10k 100k 1m 4 3 2 1 v s = 5v a vcl = 1 r l = 2k v t a = 25 8 c figure 13. maximum output swing vs. frequency frequency C hz output swing C v p-p 20 18 0 100 1k 10m 10k 100k 1m 10 6 2 v s = 18v a vcl = 1 r l = 2k v t a = 25 8 c 4 8 16 14 12 figure 14. maximum output swing vs. frequency frequency C hz impedance C v 300 240 0 1k 10k 100k 1m 10m 180 120 60 5v # v s # 18v t a = 25 8 c a v = 100 a v = 10 a v = 1 figure 15. closed-loop output impedance vs. frequency frequency C hz gain C db 40 1k 10k 100m 100k 1m 10m 20 0 5v # v s # 18v t a = 25 8 c figure 16. closed-loop gain vs. frequency frequency C hz common-mode rejection C db 100 100 1k 10m 10k 100k 1m 80 60 5v # v s # 18v t a = 25 8 c 0 20 40 120 140 figure 17. common-mode rejection vs. frequency frequency C hz power-supply rejection C db 100 1k 10k 100k 1m 100 0 80 60 40 20 10m v s = 18v t a = 25 8 c psrr+ psrr 2 figure 18. power-supply rejection vs. frequency
ad8614/ad8644 C5C rev. 0 supply voltage C v 0 220 4681012141618 slew rate C v/ m s 9 8 0 4 3 2 1 7 5 6 a v = 1 r l = 2k v c l = 200pf t a = 25 8 c sr+ sr 2 figure 19. slew rate vs. supply voltage voltage noise density C nv hz frequency C hz 100 10 1 10 100 10k 1k v s = 5v t a = 25 8 c figure 20. voltage noise density vs. frequency voltage noise density C nv hz frequency C hz 100 10 1 10 100 10k 1k v s = 18v t a = 25 8 c figure 21. voltage noise density vs. frequency applications section theory of operation the ad8614/ad8644 are processed using analog devices high voltage, high speed, complementary bipolar processhv xfcb. this process includes trench isolated transistors that lower parasitic capacitance. figure 22 shows a simplified schematic of the ad8614/ad8644. the input stage is rail-to-rail, consisting of two complementary differential pairs, one npn pair and one pnp pair. the input stage is protected against avalanche breakdown by two back-to-back diodes. each input has a 1.5 k w resistor that limits input current during over-voltage events and furnishes phase reversal protection if the inputs are exceeded. the two differential pairs are connected to a double-folded cascode. this is the stage in the amplifier with the most gain. the double folded cascode differentially feeds the output stage circuitry. two complementary common emitter tran- sistors are used as the output stage. this allows the output to swing to within 125 mv from each rail with a 10 ma load. the gain of the output stage, and thus the open loop gain of the op amp, depends on the load resistance. v cc 2 + 1.5k v v ee v cc v out 1.5k v v cc figure 22. simplified schematic the ad8614/ad8644 have no built-in short circuit protection. the short circuit limit is a function of high current roll-off of the output stage transistors and the voltage drop over the resistor shown on the schematic at the output stage. the voltage over this resistor is clamped to one diode during short circuit voltage events. output short-circuit protection to achieve a wide bandwidth and high slew rate, the output of the ad8614/ad8644 is not short-circuit protected. shorting the output directly to ground or to a supply rail may destroy the device. the typical maximum safe output current is 70 ma. in applications where some output current protection is needed, but not at the expense of reduced output voltage headroom, a low value resistor in series with the output can be used. this is shown in figure 23. the resistor is connected within the feedback loop of the amplifier so that if v out is shorted to ground and v in swings up to 18 v, the output current will not exceed 70 ma. for 18 v single supply applications, resistors less than 261 w are not recommended.
ad8614/ad8644 C6C rev. 0 ad86x4 v in 261 v v out 18v figure 23. output short-circuit protection input overvoltage protection as with any semiconductor device, whenever the condition exists for the input to exceed either supply voltage, attention needs to be paid to the input overvoltage characteristic. as an overvoltage occurs, the amplifier could be damaged, depending on the voltage level and the magnitude of the fault current. when the input voltage exceeds either supply by more than 0.6 v, internal pin junctions energize, allowing current to flow from the input to the supplies. observing figure 22, the ad8614/ad8644 has 1.5 k w resistors in series with each input, which helps limit the current. this input current is not inherently damaging to the device as long as it is limited to 5 ma or less. if the voltage is large enough to cause more than 5 ma of cur- rent to flow, an external series resistor should be added. the size of this resistor is calculated by dividing the maximum overvoltage by 5 ma and subtracting the internal 1.5 k w resistor. for example, if the input voltage could reach 100 v, the external resistor should be (100 v/5 ma) C 1.5 k w = 18.5 k w . this resistance should be placed in series with either or both inputs if they are subjected to the over- voltages. for more information on general overvoltage characteristics of amplifiers refer to the 1993 system applications guide , available from the analog devices literature center. output phase reversal the ad8614/ad8644 is immune to phase reversal as long as the input voltage is limited to within the supply rails. alt hough the devices output will not change phase, large currents due to input overvoltage could result, damaging the device. in applica- tions where the possibility of an i nput voltage exceeding the supply voltage exists, over voltage protection should be used, as described in the previous section. power dissipation the maximum power that can be safely dissipated by the ad8614/ad8644 is limited by the associated rise in junction temperature. the maximum safe junction temperature is 150 c, and should not be exceeded or device performance could suffer. if this maximum is momentarily exceeded, proper circuit opera- tion will be restored as soon as the die temperature is reduced. leaving the device in an overheated condition for an e xtended period can result in permanent damage to the device. to calculate the internal junction temperature of the ad86x4, the following formula can be used: t j = p diss q ja + t a where: t j = ad86x4 junction temperature; p diss = ad86x4 power dissipation; q ja = ad86x4 package thermal resistance, junction-to- ambient; and t a = ambient temperature of the circuit. the power dissipated by the device can be calculated as: p diss = i load ( v s C v out ) where: i load is the ad86x4 output load current; v s is the ad86x4 supply voltage; and v out is the ad86x4 output voltage. figure 24 provides a convenient way to see if the device is being overheated. the maximum safe power dissipation can be found graphically, based on the package type and the ambient tem- perature around the package. by using the previous equation, it is a simple matter to see if p diss exceeds the devices power derating curve. to ensure proper operation, it is important to observe the recommended derating curves shown in figure 24. ambient temperature C 8 c 1.5 0 C35 C15 5 25 45 65 85 1.0 0.5 14-lead soic package u ja = 120 8 c/w 14-lead tssop package u ja = 180 8 c/w 5-lead sot-23 package u ja = 230 8 c/w maximum power dissipation C watts figure 24. maximum power dissipation vs. temperature for 5-lead and 14-lead package types unused amplifiers it is recommended that any unused amplifiers in the quad pack- age be configured as a unity gain follower with a 1 k w feedback resistor connected from the inverting input to the output, and the noninverting input tied to the ground plane. capacitive load drive the ad8614/ad8644 exhibits excellent capacitive load driving capabilities. although the device is stable with large capacitive loads, there is a decrease in amplifier bandwidth as the capacitive load increases. when driving heavy capacitive loads directly from the ad8614/ ad8644 output, a snubber network can be used to improve the transient response. this network consists of a series r-c connected from the amplifiers output to ground, placing it in parallel with the capacitive load. the configuration is shown in figure 25. although this network will not increase the bandwidth of the amplifier, it will significantly reduce the amount of overshoot. ad86x4 v in v out 5v r x c x c l figure 25. snubber network compensation for capacitive loads
ad8614/ad8644 C7C rev. 0 the optimum values for the snubber network should be determined empirically based on the size of the capacitive load. table i shows a few sample snubber network values for a given load capacitance. table i. snubber networks for large capacitive loads load capacitance snubber network (c l )(r s , c s ) 0.47 nf 300 w , 0.1 m f 4.7 nf 30 w , 1 m f 47 nf 5 w , 1 m f direct access arrangement figure 26 shows a schematic for a 5 v single supply transmit/receive telephone line interface for 600 w transmission systems. it allows full duplex transmission of signals on a transformer-coupled 600 w line. amplifier a1 provides gain that can be adjusted to meet the modem output drive requirements. both a1 and a2 are configured to apply the largest possible differential signal to the transformer. the largest signal available on a single 5 v supply is approximately 4.0 v p-p into a 600 w transmission system. amplifier a3 is config- ured as a difference amplifier to extract the receive information from the transmission line for amplification by a4. a3 also prevents the transmit signal from interfering with the receive signal. the gain of a4 can be adjusted in the same manner as a1s to meet the modems input signal requirements. standard resistor values permit the use of sip (single in-line package) format resistor arrays. couple this with the ad8644 14-lead soic or tssop package and this circuit can offer a compact solution. 6.2v 6.2v transmit txa receive rxa c1 0.1 m f r1 10k v r2 9.09k v 2k v p1 tx gain adjust a1 a2 a3 a4 a1, a2 = 1/2 ad8644 a3, a4 = 1/2 ad8644 r3 360 v 1:1 t1 to telephone line 1 2 3 7 6 5 2 3 1 6 5 7 10 m f r7 10k v r8 10k v r5 10k v r6 10k v r9 10k v r14 14.3k v r10 10k v r11 10k v r12 10k v r13 10k v c2 0.1 m f p2 rx gain adjust 2k v z o 600 v 5v dc midcom 671-8005 figure 26. a single-supply direct access arrangement for modems a one-chip headphone/microphone preamplifier solution because of its high output current performance, the ad8644 makes an excellent amplifier for driving an audio output jack in a computer application. figure 27 shows how the ad8644 can be interfaced with an ac codec to drive headphones or speakers u1-a r1 2k v 4 c1 100 m f 5v 1 10 2 3 5 5v v dd v dd left out ad1881 (ac'97) right out v ss r3 20 v 7 8 6 9 r4 20 v c2 100 m f note: additional pins omitted for clarity u1-b u1 = ad8644 r2 2k v 28 35 36 figure 27. a pc-99 compliant headphone/line out amplifier if gain is required from the output amplifier, four additional resistors should be added as shown in figure 28. the gain of the ad8644 can be set as: a r r v = 6 5 u1-a r1 2k v 4 c1 100 m f 5v 1 10 2 3 5 5v v dd v dd left out ad1881 (ac97) right out v ss r3 20 v 7 8 6 9 r4 20 v c2 100 m f note: additional pins omitted for clarity u1-b u1 = ad8644 r2 2k v r6 20k v r6 20k v v ref r5 10k v r5 10k v a v = = +6db with values shown r6 r5 38 35 27 36 figure 28. a pc-99-compliant headphone/speaker amplifier with gain input coupling capacitors are not required for either circuit as the reference voltage is supplied from the ad1881. r4 and r5 help protect the ad8644 output in case the output jack or headphone wires are accidentally shorted to ground. the output coupling capacitors c1 and c2 block dc current from the headphones and create a high-pass filter with a corner frequency of: f cr r db l - = + () 3 1 214 p where r l is the resistance of the headphones.
C8C rev. 0 c3735C8C10/99 printed in u.s.a. ad8614/ad8644 outline dimensions dimensions shown in inches and (mm). 5-lead sot-23 (rt suffix) 0.1181 (3.00) 0.1102 (2.80) pin 1 0.0669 (1.70) 0.0590 (1.50) 0.1181 (3.00) 0.1024 (2.60) 1 3 4 5 0.0748 (1.90) bsc 0.0374 (0.95) bsc 2 0.0079 (0.20) 0.0031 (0.08) 0.0217 (0.55) 0.0138 (0.35) 10 8 0 8 0.0197 (0.50) 0.0138 (0.35) 0.0059 (0.15) 0.0019 (0.05) 0.0512 (1.30) 0.0354 (0.90) seating plane 0.0571 (1.45) 0.0374 (0.95) 14-lead tssop (ru suffix) 14 8 7 1 0.201 (5.10) 0.193 (4.90) 0.256 (6.50) 0.246 (6.25) 0.177 (4.50) 0.169 (4.30) pin 1 seating plane 0.006 (0.15) 0.002 (0.05) 0.0118 (0.30) 0.0075 (0.19) 0.0256 (0.65) bsc 0.0433 (1.10) max 0.0079 (0.20) 0.0035 (0.090) 0.028 (0.70) 0.020 (0.50) 8 8 0 8 14-lead narrow soic (r suffix) 14 8 7 1 0.3444 (8.75) 0.3367 (8.55) 0.2440 (6.20) 0.2284 (5.80) 0.1574 (4.00) 0.1497 (3.80) pin 1 seating plane 0.0098 (0.25) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.0688 (1.75) 0.0532 (1.35) 0.0500 (1.27) bsc 0.0099 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 8 8 0 8 0.0196 (0.50) 0.0099 (0.25) x 45 8 the remaining two amplifiers can be used as low voltage microphone preamplifiers. a single ad8614 can be used as a stand-alone microphone preamplifier. figure 29 shows this implementation. ad1881 (ac'97) v ref 10k v 21 mic 1 in a v = 20db 1k v 1 m f 2.2k v 5v 10k v a v = +20db 1k v 1 m f 2.2k v 5v mic 1 mic 2 22 mic 2 in 27 figure 29. microphone preamplifier spice model availability the spice model for the ad8614/ad8644 amplifier is available and can be downloaded from the analog devices web site at http://www.analog.com . the macro-model accurately simulates a number of ad8614/ad8644 parameters, including offset volt- age, input common-mode range, and rail-to-rail output swing. the output voltage versus output current characteristic of the macro-model is identical to the actual ad8614/ad8644 perfor- mance, which is a critical feature with a rail-to-rail amplifier model. the model also accurately simulates many ac effects, such as gain bandwidth product, phase margin, input voltage noise, cmrr and psrr versus frequency, and transient response. its high degree of model accuracy makes the ad8614/ad8644 macro-model one of the most reliable and true-to-life models available for any amplifier.
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